Method for using a non-orthogonal pilot signal with data channel interference cancellation

ABSTRACT

A system and method for encoding/decoding data channels in a CDMA system having data channel interference cancellation, wherein data channel interference cancellation is used to remove unwanted non-orthogonal pilot signal components which are present within a demodulated data signal. This is accomplished by regenerating interference terms with respect to the non-orthogonal pilot signal and subtracting them from the demodulated data signal.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No. 11/475,447, filed Jun. 27, 2006, which is a continuation of U.S. application Ser. No. 11/178,767, filed Jul. 11, 2005, which is a continuation of U.S. application Ser. No. 09/772,200 filed Jan. 29, 2001, which claims the benefit of U.S. Provisional Application No. 60/184,365 filed on Feb. 23, 2000, which are incorporated by reference as if fully set forth.

FIELD OF INVENTION

The present invention generally relates to the field of data communications and processing and, more particularly, to a method for encoding/decoding data channels in a CDMA system having data channel interference cancellation.

BACKGROUND

Code Division Multiple Access (CDMA) modulation is a multi-user access transmission scheme in which different users of the same transmission medium overlap both in frequency and in time. This is in contrast to Frequency Division Multiple Access (FDMA) in which users overlap in time, but are assigned unique frequencies, and Time Division Multiple Access (TDMA) in which users overlap in frequency, but are assigned unique timeslots. According to CDMA, each user is assigned a unique code sequence that allows the user to spread its information over the entire channel bandwidth, as opposed to particular sub-channel(s) in FDMA. Thus, signals from all users are transmitted over the entire channel. To separate out the signals for a particular user at a receiver, cross correlation is performed on the received signal using the same unique user code sequence.

CDMA transmission is well known to those of skill in the art. A comparison between CDMA and FDMA/TDMA may be found in Proakis, “Digital Communications”, Chapter 15, which is incorporated herein by reference. Also, an example of a combined approach for minimizing inter-user interference (i.e., combining a Walsh basis within a group and a spreading sequence across groups) is the IS-95 system described in TIA/EIA/IS-95 “Mobile Station Compatibility Standard for Dual Mode Wideband Spread Spectrum Cellular System”, which is incorporated herein by reference.

An IS-95 CDMA system is unique in that its forward and reverse links (i.e., the base station to mobile station and mobile station to base station) have different link structures. This is necessary to accommodate the requirements of a land-mobile communication system. The forward link consists of four types of logical channels, i.e., pilot, sync, paging, and traffic channels, with one pilot channel, one sync channel, up to seven paging channels, and several traffic channels. Each of these forward-linked channels is first spread orthogonally by its Walsh function, and then spread by a pair of short PN sequences (so-called pseudonoise) each of which is a sequence of high data rate bits (“Chips”) ranging from −1 to +1 (polar) or 0 to 1 (non-polar). Subsequently, all channels in the system are added together to form the composite spread spectrum signal which is transmitted on the forward link.

The reverse link in the IS-95 CDMA system consists of two types of logical channels, i.e., access and traffic channels. Each of these reverse-link channels is spread orthogonally by a unique long PN sequence; hence each channel is recovered or decoded using the distinct long PN code. In some instances, a pilot channel is not used on the reverse link based on the impracticality of each mobile station broadcasting its own pilot sequence. Additionally, the IS-95 CDMA system uses 64 Walsh functions which are orthogonal to each other (i.e., their cross-product is equal to zero), and each of the logic channels on the forward link is identified by its assigned Walsh function. The Walsh function is used to generate a code which is used to separate individual users occupying the same RF band to avoid mutual interference on the forward link. The access channel is used by the mobile station to communicate with the base station when a traffic channel is not assigned to the mobile station. The mobile station uses the access channel to generate call originations and respond to pages and orders. The baseband data rate of the access channel is fixed at 4.8 kilobits per second (Kbps).

The pilot channel is identified by the Walsh function 0 (ω₀). This channel contains no baseband sequence information. The baseband sequence is a stream of 0s which are spread by Walsh function 0, which is also a sequence of all zeros. The resulting sequence (still all 0s) is then spread or multiplied by a pair of quadrature PN sequences. Therefore, the pilot channel is effectively the PN sequence itself. The PN sequence with a specified offset uniquely identifies the particular geographical area or sector from which the user is transmitting the pilot signal. In an IS-95 CDMA system, both Walsh function 0 and the PN sequence operate at a rate of 1.2288 mega chips per second (Mcps). After PN spreading, baseband filters are used to shape the resultant digital pulses. These filters effectively lowpass filter the digital pulse stream and control the baseband spectrum of the signal. As a result, the signal band possesses a sharper roll-off near the band edge. The pilot channel is transmitted continuously by the base station sector. The pilot channel provides the mobile station with timing and phase reference. The measurement of the signal-to-noise ratio of the pilot channel by the mobile station also provides an indication of the strongest serving sector of that mobile. Here, the signal-to noise is the energy per chip per interference density, or E_(c)/I₀ where E_(c) is the energy per chip and I₀ is the interference density.

Unlike the pilot channel, the sync channel carries baseband information. The baseband information is contained in the sync channel message which notifies the mobile of information concerning system synchronization and parameters. Similar to the sync channel, the paging channel also carries baseband information. However, unlike the sync channel, the paging channel transmits at a higher rate, i.e., at either 4.8 or 9.6 Kbps.

The forward and reverse traffic channels are used to transmit user data and voice; signaling messages are also sent over the traffic channel. The structure of the forward traffic channel is similar to that of the paging channel, while the structure of the reverse traffic channel is similar to that of the access channel. The only difference is that the forward traffic channel contains multiplexed power control bits (PCBs) and the reverse traffic channel contains a data burst randomizer which is used to generate a masking pattern of 0s and 1s to randomly mask redundant data.

The techniques for separating signals in time (i.e., TDMA), or in frequency (i.e., FDMA) are relatively simple ways of ensuring that the signals are orthogonal and noninterfereing. However, in CDMA, different users occupy the same bandwidth at the same, but are separated from each other via the use of a set of orthogonal waveforms, sequences, or codes. Two real-valued waveforms x and y are said to be orthogonal if their cross correlation R_(xy) over time period T is zero, where

$\begin{matrix} {{R_{xy}(0)} = {\int_{0}^{T}{{x(t)}{y(t)}\ {t}}}} & \left( {{Eq}.\mspace{14mu} 1} \right) \end{matrix}$

In discrete time, the two sequences x and y are orthogonal if their cross-product R_(xy)(0) is zero. The cross product is defined as

$\begin{matrix} {{{R_{xy}(0)} = {{x^{T}y^{T}} = {\sum\limits_{i = 1}^{I}\; {x_{i}y_{i}}}}}{where}{x^{T} = \left\lbrack {x_{1}\mspace{11mu} x_{2}\mspace{11mu} \ldots \mspace{14mu} x_{i}} \right\rbrack}{y^{T} = \left\lbrack {y_{1}\mspace{11mu} y_{2}\mspace{11mu} \ldots \mspace{14mu} y_{i}} \right\rbrack}} & \left( {{Eq}.\mspace{14mu} 2} \right) \end{matrix}$

In this case, T denotes the vector transpose, i.e., a column represented as a row or vice versa. For example, the following two sequences or codes, x and y are orthogonal:

x ^(T)=[−1−111]

y ^(T)=[−111−1]

because their cross-correlation is zero; that is

R _(xy)(0)=x ^(T) y ^(T)=(−1)(−1)+(−1)(1)+(1)(1)+(1)(−1)  (Eq. 3)

In order for the set of codes to be used in a multiple access scheme, additional properties are required. That is, in addition to the zero cross-correlation property, each code in the set of orthogonal codes must have an equal number of 1s and −1s. This property provides each particular code with the required pseudorandom characteristic. An additional property is that the dot product of each code scaled by the order of the code must equal to 1. The order of the code is effectively the length of the code, and the dot product is defined as a scalar obtained by multiplying the sequence by itself and summing the individual terms. This is given by the following relationship:

$\begin{matrix} {{R_{xx}(0)} = {{x^{T}x} = {\sum\limits_{i = 1}^{I}\; {x_{i}x_{i}}}}} & \left( {{Eq}.\mspace{14mu} 4} \right) \end{matrix}$

The increasing use of wireless telephones and personal computers has led to a corresponding demand for such advanced telecommunications techniques as CDMA, FDMA and TDMA, which were once thought to be only meant for use in specialized applications. In the 1980's wireless voice communication became widely available through the cellular telephone network. Such services were at first typically considered to be the exclusive province of the businessman because of high subscriber costs. The same was also true for access to remotely distributed computer networks, whereby until very recently, only business people and large institutions could afford the necessary computers and wireline access equipment. As a result of the widespread availability of both technologies, the general population now increasingly wishes to not only have access to networks such as the Internet and private intranets, but also to access such networks in a wireless manner as well. This is of particular concern to the users of portable computers, laptop computers, hand-held personal digital assistants and the like who prefer to access such networks without being tethered to a telephone line.

However, there is still no widely available satisfactory solution for providing low cost, broad geographical coverage, high speed access to the Internet, private intranets, and other networks using the existing wireless infrastructure. This situation is a result of several factors. For one, the typical manner of providing high speed data service in the business environment over the wireline network is not readily adaptable to the voice grade service which is available in most homes or offices. Additionally, such standard high speed data services do not lend themselves well to efficient transmission over standard cellular wireless handsets. Furthermore, the existing cellular network was originally designed only to deliver voice services. As a result, the emphasis in present day digital wireless communication schemes lies with voice, although certain schemes such as CDMA do provide some measure of asymmetrical behavior for the accommodation of data transmission. For example, the data rate on an IS-95 forward traffic channel can be adjusted in increments from 1.2 Kbps to up to 9.6 Kbps for so-called Rate Set 1, and for increments from 1.8 Kbps up to 14.4 Kbps for Rate Set 2.

Existing systems therefore typically provide a radio channel which can accommodate maximum data rates only in the range of 14.4 Kbps at best in the forward direction. Such a low rate data channel does not directly lend itself to transmitting data at rates of 28.8 or even 56.6 Kbps which are now commonly available with conventional modem type equipment. Data rates at these levels are rapidly becoming the minimum acceptable rates for activities such as Internet access. Other types of data networks using higher speed building blocks such as Digital Subscriber Line (xDSL) service are just now coming into use. However, the cost of xDSL service has only recently been reduced to the point where it is attractive to the residential customer.

Although xDSL and Integrate Services Digital Network (ISDN) networks were known at the time that cellular systems were originally deployed, for the most part, there is no provision for providing higher speed ISDN or xDSL grade data services over cellular networks. Unfortunately, in wireless environments, access to channels by multiple subscribers is expensive and there is competition for them. Whether the multiple access is provided by the traditional FDMA using analog modulation on a group of radio carriers, or by the newer digital modulation schemes which permit sharing of a radio carrier using TDMA or CDMA, the nature of the radio spectrum is that it is a medium which is expected to be shared. This is quite different from the traditional environment for data transmission, in which the wireline medium is relatively inexpensive to obtain, and is therefore not typically intended to be shared. Accordingly, it is apparent that there is a need to provide a system which supports higher speed ISDN or xDSL grade data services over cellular network topologies. In particular, what is needed is an efficient scheme for supporting wireless data communication such as from portable computers to computer networks such as the Internet and private intranets using widely available infrastructure.

Most modern wireless standards in widespread use such as CDMA do not provide an adequate structure with which to support the most common activities, such as web page browsing. In the forward and reverse link direction, the maximum available channel bandwidth in an IS-95 type CDMA system is only 14.4 Kbps. Due to IS-95 being circuit-switched, there are only a maximum of 64 circuit-switched users that can be active at one time. In practicality, this limit is difficult to attain, and 20 or 30 simultaneous users are typically active at one time. Furthermore, existing CDMA systems require certain operations before a channel can be used. For example, both access and traffic channels are modulated by so-called long code pseudonoise (PN) sequences. In addition, in order for the receiver to work properly it must first be synchronized with the transmitter. The setting up and tearing down of user channels therefore requires overhead to perform such synchronization. This overhead results in a reduction of the system data rate which produces a noticeable delay to a user of a subscriber unit. Moreover, in the presence of benign cell conditions, the data rate of a conventional CDMA system may become limited by the number of available orthogonal code channels.

SUMMARY

The present invention is directed to a method for encoding/decoding data channels in a system having data channel interference cancellation. In accordance with the invention, the data rate of a system for a given user is increased by using a non-orthogonal pilot signal for channelization. As a result, one or more orthogonal channels become available for user traffic, rather than for use by the pilot channel. This leads to a reduction in the number of occupied orthogonal channels and an increase in system capacity available for each user due to the attainment of higher data rates which permit faster data delivery to system subscribers.

The use of a non-orthogonal pilot signal requires interference cancellation to remove the modulation effects of the pilot signal upon the data signal. This is accomplished by regenerating interference terms with respect to the non-orthogonal pilot signal and subtracting them from the demodulated data signal.

BRIEF DESCRIPTION OF THE DRAWINGS

A more detailed understanding may be had from the following description, given by way of example in conjunction with the accompanying drawings wherein:

FIG. 1 is a block diagram of a wireless communication system which uses interference cancellation on the pilot channel in accordance with the invention;

FIG. 2 is a schematic block diagram of a CDMA transceiver for implementing the method in accordance with the present invention;

FIG. 3 is an illustration of a pilot/data spreader of FIG. 2;

FIG. 4 is an illustration of a data despreader of FIG. 2;

FIG. 5 is an illustration of a pilot despreader of FIG. 2;

FIG. 6 is an illustration of an interference cancellor of FIG. 2;

FIG. 7 is an illustration of a dot product calculator of FIG. 2; and

FIGS. 8A and 8B are flow charts illustrating the steps of the method according to the invention.

DETAILED DESCRIPTION

When referred to hereafter, the terminology “wireless transmit/receive unit (WTRU)” includes but is not limited to a user equipment (UE), a mobile station, a fixed or mobile subscriber unit, a pager, a cellular telephone, a personal digital assistant (PDA), a computer, or any other type of user device capable of operating in a wireless environment. When referred to hereafter, the terminology “base station” includes but is not limited to a Node-B, a site controller, an access point (AP), or any other type of interfacing device capable of operating in a wireless environment.

FIG. 1 is a block diagram of a wireless communication system 100 which uses data channel interference cancellation to remove unwanted non-orthogonal pilot signal components present within the data signal. This results in a reduction in the number of occupied orthogonal channels and an increase in system capacity. This yields an increase in the system data rate which results in a reduction of the delay experienced by the user of the subscriber unit. As a result, high speed data and voice service over a wireless connection is achieved.

The system 100 includes two different types of components, such as subscriber units 101-1, 101-2, . . . , 101-u (collectively, the subscriber unit 101) and one or more base stations 170. The subscriber units 101 and base stations 170 cooperate to provide the functions necessary to provide wireless data services to a portable computing device 110 such as a laptop computer, portable computer, personal digital assistance (PDA) or the like associated with a subscriber unit 101. The base station 170 also cooperates with the subscriber units 101 to permit the ultimate transmission of data to and from the subscriber unit 101 and the public switch telephone network (PSTN) 180. More particularly, data and/or voice services are also provided by the subscriber unit 101 to the portable computer 110 as well as one or more devices such as telephones. The telephones themselves may in turn be connected to other modems and computers which are not shown in FIG. 1.

The subscriber unit 101 itself may include a modem, such as an ISDN modem 120, a device referred to herein as a protocol converter 130 which performs various functions including spooling 132 and bandwidth management 134, CDMA transceiver 140, and subscriber unit antenna 150. The various components of the subscriber unit 101 may be realized in discrete devices or as an integrated unit. For example, an existing conventional ISDN modem 120 such as is readily available from any number of manufacturers may be used together with existing CDMA transceivers 140. In this case, the necessary additional functions may be provided entirely by the protocol converter 130 which may be sold as a separate device. Alternatively, the ISDN modem 120, protocol converter 130 and CDMA transceiver 140 may be integrated as a complete unit and sold as a single subscriber unit device 101. Other types of interface connections such as Ethernet or PCMCIA may be used to connect the computing device to the protocol converter 130. The device may also interface to an Ethernet interface rather than an ISDN “U” interface.

The ISDN modem 120 converts data and voice signals between the format used by the terminal equipment 110 and the format required by the standard ISDN “U” interface. The U interface is a reference point in ISDN systems that designates a point of the connection between the network termination (NT) and the telephone company.

The protocol converter 130 performs spooling 132 and basic bandwidth management 134 functions. In general, spooling 132 consists of insuring that the subscriber unit 101 communicates with the terminal equipment 110 which is connected to the public switched telephone network 180 on the other side of the base station 170 at all times. The bandwidth management function 134 is responsible for allocating and deallocating CDMA radio channels 160 as required. Bandwidth management 134 also includes the dynamic management of the bandwidth allocated to a given session by dynamically assigning sub-portions of the CDMA radio channels 160. The CDMA transceiver 140 accepts the data from the protocol converter 130 and reformats the data into the appropriate form for transmission through the subscriber unit antenna 150 over CDMA radio link 160-1. The CDMA transceiver 140 may operate over only a single 1.25 MHz radio frequency channel, or may be tunable over multiple allocatable radio frequency channels.

CDMA signal transmissions from the subscriber units 101 are received and processed by the base station equipment 170. The base station equipment 170 typically includes multichannel antennas 171, multiple CDMA transceivers 172 and a bandwidth management function 174. Bandwidth management 174 controls the allocation of CDMA radio channels 160 and subchannels, in a manner analogous to the subscriber unit 101. Transceiver 172 demodulates the received CDMA signals, and the base station 170 couples the demodulated radio signals to the PSTN 180 in a manner which is well known in the art. For example, the base station 170 may communicate with the PSTN 180 over any number of different efficient communication protocols such as primary rate ISDN, or other LAPD based protocol such as IS-634 or V5.2.

It should also be understood that data signals travel bidirectionally across the CDMA radio channels 160. In other words, data signals received from the PSTN 180 are coupled to the portable computer 110 in a forward link direction, and data signals originating at the portable computer 110 are coupled to the PSTN 180 in a reverse link direction.

Each of the CDMA transceivers such as transceiver 140 in the subscriber unit 101, and transceivers 172 in the base station 170, are capable of being tuned at any given point in time to a given 1.25 Megahertz radio frequency channel. It is generally understood that such 1.25 MHz radio frequency carrier provides, at best, a total equivalent of about 500, 600 kbps maximum data rate transmission within acceptable bit error rate limitations.

FIG. 2 is a schematic block diagram of CDMA transceivers 140, 172 of the wireless communication system 100 for implementing the method according to the present invention. Specifically, FIG. 2 is a block diagram of a transmitter portion of a transceiver 140 and a receiver portion of transceiver 172. Initially, pilot spreader 201 is used to modulate a non-orthogonal pilot signal such that the pilot signal is spread over an entire channel bandwidth. Concurrently, data spreader 204 is used to spread data over the same channel bandwidth. The spread pilot and data signals are then combined to form a composite signal S(t) which is transmitted to base station 170 for despreading by pilot despreader 202 and data despreader 205, respectively. The despreaders 202, 205 are used to recover the non-orthogonal pilot signal and the data signal, respectively, from the transmitted composite signal S(t). The outputs of the pilot despreader 202 and data despreader 205 are fed to an interference canceller 203 which is used to remove interference introduced into the data signal by the non-orthogonal pilot signal. Once the interference from the non-orthogonal pilot signal is removed by the interference canceller 203, the original data is recovered via dot product calculator 206 and output for later processing by a communications system (not shown).

FIG. 3 is a block diagram of a pilot/data spreader 201 and 204 of FIG. 2 which are used to modulate the non-orthogonal pilot and data signals such that they are spread over an entire channel bandwidth. At nodes 201 a and 201 b of the pilot spreader 201, a non-orthogonal pilot signal P is modulated by a channel code p_(c), which is used to uniquely identify the transmitted pilot signal P. At nodes 204 a and 204 b of the data spreader 204, a data signal which is split into sub-band data I and Q is mixed with a signal g_(i) which represents a specific channel code of a user (I and Q represent the in-phase and quadrature portions of the data signal, respectively). At node 201 c/204 c, the output signal from node 201 a is summed with the output signal from node 204 a to produce a resultant signal. Simultaneously, at node 201 d/204 d, the output signal from node 204 b is summed with the output signal from node 201 b to produce a resultant signal.

At nodes 201 e/204 e and 201 f/204 f, the resultant signals are each modulated by a PN code a. Next, in order to provide baseband or phase discrimination between the I and Q sub-band portions of the data signal, the output signals of nodes 201 e/204 e and 201 f/204 f are modulated (i.e., spread) by channel separation signals w_(I) and w_(Q), respectively, at nodes 201 g/204 g and 201 h/204 h, respectively. In this case, the channel separation signals w_(I) and w_(Q) belong to a family of orthogonal functions such as those disclosed in U.S. Pat. No. 4,460,992 to Gutleber, which is incorporated herein by reference as if set forth expressly. Each respective channel separation signal spreads the in-phase portion and quadrature portion of the data signal to produce composite signals. At the nodes 201 i/204 i and 201 j/204 j, the respective composite output signals from nodes 201 g/204 g and 201 h/204 h are subsequently modulated by respective cosine and sine functions (i.e., cos(wt+θ) and sin(wt+θ)). The output signals from nodes 201 i/204 i and 204 j/201 j are then summed to form a composite signal S(t) given by the following relationship:

S(t)=Pap _(c) w ₁ cos(wt+θ)+Pap _(c) w _(Q) sin(wt+θ)+I _(n) aw _(I) g _(i) cos(wt+θ)+Q _(n) aw _(Q) g _(i) sin(wt+θ)  (Eq. 5)

The signal given by the relationship in equation 5 is transmitted to base station 170 which contains a data despreader 205 (see FIG. 6) for use in the demodulation of the transmitted composite signal S(t) to recover the original data signal.

FIG. 4 is a schematic block diagram of a data despreader 205 which is used in the recovery of the originally transmitted data signal. In the data despreader 205 shown in FIG. 4, the signal S(t) given in equation 5 is initially decoded by demodulating S(t) by cos(wt) and sin(wt) at nodes 205 a and 205 b, respectively to produce resultant output signals. Next, at nodes 205 c and 205 d, the resultant output signals from nodes 205 a and 205 b are demodulated by the PN code a. The output signals of nodes 205 c and 205 d are each demodulated by the channel separation function w_(Q) at nodes 205 f and 205 g, respectively. Concurrently, the output signal of node 205 c is demodulated by the channel separation function w, at node 205 e, while at node 205 h the output signal of node 205 d is demodulated by a channel separation function −w_(I) which is a complex conjugate of the channel separation function w_(I). The output signals of nodes 205 e, 205 f, 205 g and 205 h are respectively demodulated at nodes 205 i, 205 j, 205 k and 205 l by the channel code of a user g_(i).

Given two codes A and B of length n, an integration and dump function occurs when the lengths of the codes are matched, multiplied together, integrated and the result output for further processing. In this manner, an integration and dump function is then performed at nodes 205 m-205 p, respectively, upon the output signals of nodes 205 i-205 l to obtain the following relationships:

$\begin{matrix} {{\sum\limits_{N}\; {\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{Q}g_{i}}} = {{\frac{N}{2}Q_{n}\mspace{11mu} {\sin (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}\mspace{11mu} {\sin (\theta)}}}} + {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}w_{I}w_{Q}\mspace{11mu} {\cos (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 6} \right) \\ {{\sum\limits_{N}\; {\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{I}g_{i}}} = {{\frac{N}{2}I_{n}\mspace{11mu} {\cos (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}\mspace{11mu} {\cos (\theta)}}}} + {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}w_{I}w_{Q}\mspace{11mu} {\sin (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 7} \right) \\ {{\sum\limits_{N}\; {\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){a\left( {- w_{i}} \right)}g_{i}}} = {{\frac{N}{2}I_{n}\mspace{11mu} {\sin (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}\mspace{11mu} {\sin (\theta)}}}} - {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}w_{I}w_{Q}\mspace{11mu} {\cos (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 8} \right) \\ {{\sum\limits_{N}\; {\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){aw}_{Q}g_{i}}} = {{\frac{N}{2}Q_{n}\mspace{11mu} {\cos (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}\mspace{11mu} {\cos (\theta)}}}} - {\frac{1}{2}{\sum\limits_{N}\; {{Pg}_{i}p_{c}w_{I}w_{Q}\mspace{11mu} {\sin (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 9} \right) \end{matrix}$

where each summation term in equations 6-9 represents interference due to the pilot signal which must be removed to accurately reconstruct the originally transmitted data signal, and each N in the summation is the processing gain.

FIG. 5 is an illustration of the pilot despreader 202 which is used to recover the originally transmitted pilot signal P. To accomplish this, the transmitted composite signal S(t), given by the relationship in equation 14, is demodulated by cosine and sine functions (i.e., cos(ωt) and sin(ωt)) at nodes 202 a and 202 b. Next, the output signals from nodes 202 a and 202 b are demodulated by the PN code a at nodes 202 c and 202 d, respectively. The output signals from nodes 202 c and 202 d are each demodulated by the channel separation function w_(Q) at nodes 202 f and 202 g, respectively. Concurrently, the output signal of node 205 c is demodulated by the channel separation function w_(I), at node 202 e, while at node 202 h the output signal of node 202 d is demodulated by the channel separation function −w_(I). The output signals of nodes 202 e, 202 f, 202 g and 202 h are respectively demodulated at nodes 202 i, 202 j, 202 k and 202 l by the channel code p_(c) which is used to uniquely identify the transmitted pilot signal P. After demodulating the output signals of nodes 202 i-202 l, the integration and dump function is performed to obtain the output signals given by the following relationships at nodes 205 m, 205 n, 205 o and 205 p, respectively

$\begin{matrix} {{\sum\limits_{N}\; {\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{I}p_{c}}} = {{\frac{N}{2}P\mspace{14mu} {\cos (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {I_{n}g_{i}p_{c}\mspace{11mu} \cos \mspace{14mu} \theta}}} + {\frac{1}{2}{\sum\limits_{N}\; {Q_{n}g_{i}p_{c}w_{i}w_{Q}\mspace{11mu} {\sin (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 10} \right) \\ {{\sum\limits_{N}\; {\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{Q}p_{c}}} = {{\frac{N}{2}P\mspace{14mu} {\sin (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {Q_{n}g_{i}p_{c}\mspace{11mu} \sin \mspace{14mu} \theta}}} + {\frac{1}{2}{\sum\limits_{N}\; {I_{n}g_{i}p_{c}w_{i}w_{Q}\mspace{11mu} {\cos (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 11} \right) \\ {{\sum\limits_{N}\; {\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){a\left( {- w_{I}} \right)}p_{c}}} = {{\frac{N}{2}P\mspace{14mu} {\sin (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {I_{n}g_{i}p_{c}\mspace{11mu} \sin \mspace{14mu} \theta}}} - {\frac{1}{2}{\sum\limits_{N}\; {Q_{n}g_{i}p_{c}w_{i}w_{Q}\mspace{11mu} {\cos (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 12} \right) \\ {{\sum\limits_{N}\; {\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){aw}_{Q}p_{c}}} = {{\frac{N}{2}P\mspace{14mu} {\cos (\theta)}} + {\frac{1}{2}{\sum\limits_{N}\; {Q_{n}g_{i}p_{c}\mspace{11mu} \cos \mspace{14mu} \theta}}} - {\frac{1}{2}{\sum\limits_{N}\; {I_{n}g_{i}p_{c}w_{i}w_{Q}\mspace{11mu} {\sin (\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 13} \right) \end{matrix}$

where N is the processing gain.

As shown in FIG. 5, four output signals are generated which each contain interference as a result of the demodulation process. Equations 10-13 represent the output signal at nodes 202 m, 202 n, 202 o and 202 p, respectively. In this case, the eight summation terms in equations 10-13 represent the interference added to the pilot signal as a result of the demodulation process. At node 202 q, the output signals from nodes 202 m and 202 o are subsequently subjected to an additional integration and dump function, while the integration and dump function is performed on the output signals from nodes 202 n and 202 p at node 202 r. As a result, the signals are filtered such that the interference is removed and the originally transmitted pilot signal P is recovered.

Along with the output of the pilot despreader 202, the output of the data despreader 205 is provided to an interference canceller 203 shown in FIG. 6. The output of the pilot despreader is fed to the input of the interference canceller 203, and the output of the interference canceller 203 is subtracted from the output of the data despreader 205 in a manner which is known to yield I and Q sub-band data signals which do not contain interference associated with the pilot signal P.

The interference canceller 203 shown in FIG. 6 is used to remove the interference associated with the pilot signal P which is introduced into the data signal during the demodulation process. The interference added to the data signal is represented by the summation terms in the relationships given in equations 6-9. To remove the interference from the despread data signals, the P cos(θ) and P sin(θ) inputs of the interference canceller 203 are each modulated by the channel code p_(c) at nodes 203 a and 203 b, respectively. Next, the output signals of nodes 203 a and 203 b are each modulated by the group user channel code g_(i). At this point, an integration of the output signals of nodes 203 c and 203 d is performed to yield respective first and second interference terms given by the following relationships:

$\begin{matrix} {{\frac{P\mspace{14mu} {\cos (\theta)}}{2}{\sum\limits_{N}\; {g_{i}p_{c}}}}{and}} & \left( {{Eq}.\mspace{14mu} 14} \right) \\ {\frac{P\mspace{14mu} {\sin (\theta)}}{2}{\sum\limits_{N}\; {g_{i}p_{c}}}} & \left( {{Eq}.\mspace{14mu} 15} \right) \end{matrix}$

where N is the processing gain.

Next, the output signals from nodes 203 c and 203 d are modulated by the w, channel separation function at nodes 203 g and 203 h, respectively. The output signals from nodes 203 g and 203 h are then modulated by the channel separation function w_(Q) at nodes 203 i and 203 j, respectively. An integration of the output signals from nodes 203 i and 203 j is performed at nodes 203 k and 203 l to yield respective third and fourth interference terms given by the following relationships:

$\begin{matrix} {{\frac{P\mspace{14mu} {\cos (\theta)}}{2}{\sum\limits_{N}\; {g_{i}p_{c}w_{I}w_{Q}}}}{and}} & \left( {{Eq}.\mspace{14mu} 16} \right) \\ {\frac{P\mspace{14mu} {\sin (\theta)}}{2}{\sum\limits_{N}\; {g_{i}p_{c}w_{I}w_{Q}}}} & \left( {{Eq}.\mspace{14mu} 17} \right) \end{matrix}$

where N is the processing gain.

The relationships expressed in equations 14-17 are subtracted from the respective expressions found in equations 6-9 to remove the interference from the I and Q sub-band data signals. At this point, once the interference is removed from the data signal, complete recovery of the data signal is possible.

FIG. 7 is an illustration of an exemplary dot product calculator 206 for performing a dot product calculation to recover the original data signal. After removal of the interference terms given in equations 14-17, each respective portion of the I_(n) and Q_(n) sub-band data signals is forwarded to the dot product calculator 206. The respective cosine and sine portions of the pilot signal P which are output from the pilot despreader 202 are also forwarded to the dot product calculator 206, as shown in FIG. 7. At nodes 206 a and 206 b, the cosine portion of the pilot signal P is modulated by the cosine portions of the I_(n) and Q_(n) sub-band data signals. Simultaneously, at nodes 206 c and 206 d, the sine portion of the pilot signal P is modulated by the sine portions of the I_(n), and Q_(n) sub-band data signals. At node 206 e, the output signal of nodes 206 a and 206 c are summed together to yield an output signal given by the following relationship:

PI _(n) cos(θ−{circumflex over (θ)})≈PI _(n)  (Eq. 18)

At node 206 f, the output signals of nodes 206 b and 206 d are summed together to yield another output signal given by the following relationship:

PQ _(n) cos(θ−{circumflex over (θ)})≈PQ _(n)  (Eq. 19)

where the ̂ term in equations 18 and 19 indicates a coarse estimate of the phase over one symbol (i.e., the number of chips per signal). At this point, one skilled in the art will readily appreciate that equations 18 and 19 represent the originally transmitted I and Q sub-band data signals, where each sub-band is multiplied by the pilot signal P.

FIGS. 8A and 8B are flow charts of the method for using a non-orthogonal pilot signal according to the invention. In step 10, a non-orthogonal pilot signal P is modulated by a channel code p_(c). Simultaneously, a data signal which is split into sub-band data I and Q is mixed with a specific channel code of a user g_(i). In step 20, the non-orthogonal pilot signal is then summed with the I and Q sub-band data signals to produce resultant signals.

In step 30, the resultant signals are then modulated by a PN code a. In step 40, to provide baseband or phase discrimination between the I and Q sub-band portions of the data signal, the resultant output signals are modulated (i.e., spread) by channel separation signals w_(I), and w_(Q). In step 50, the respective composite output signals are modulated by respective cosine and sine functions (i.e., cos(wt+θ) and sin(wt+θ). In step 60, the cosine and sine output signals are then summed to form the composite signal S(t) which is transmitted to the base station 170.

In step 70, the composite signal S(t) is initially decoded by demodulating it with cos(wt) and sin(wt). Next in step 80, the resultant output signal is demodulated by the PN code a. In step 90, the resultant signal is demodulated by the channel separation function w_(Q). Concurrently, the resultant signal with respect to cos(wt) is demodulated by the channel separation function w_(I), while the resultant signal with respect to sin(wt) is demodulated by a channel separation function −w_(I). In step 100, the signals which were demodulated by the channel separation function w_(Q) are then demodulated by the channel code of a user g_(i) and the channel code p_(c).

In step 110, an integration and dump function is performed upon the resultant output signal to obtain the demodulated data signal containing the interference. Concurrently, an integration and dump function is also performed to obtain the demodulated non-orthogonal pilot signal. In step 120, the demodulated non-orthogonal pilot signal is subjected to an additional integration and dump function to remove interference from the originally transmitted non-orthogonal pilot signal P.

In step 130, the demodulated non-orthogonal pilot signal is modulated by the channel code p_(c). In step 140, the modulated pilot signal is modulated by the group user channel code g_(i). In step 150, an integration of the signal is performed to yield first and second interference terms. In step 160, the signal modulated by the user channel code g_(i) is additionally modulated by the w_(I) channel separation function. In step 170, the resultant signal is then modulated by the channel separation function w_(Q). In step 180, an integration of the resultant signal is performed to yield third and forth interference terms. In step 190, the interference terms are subtracted from the demodulated data signal to remove the interference from the I and Q sub-band data. Finally, in step 200, a dot product calculation is performed to recover the originally transmitted I and Q sub-band data signals.

While the invention has been particularly shown and described with reference to a preferred embodiment thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention.

Although features and elements are described above in particular combinations, each feature or element can be used alone without the other features and elements or in various combinations with or without other features and elements. The methods or flow charts provided herein may be implemented in a computer program, software, or firmware incorporated in a computer-readable storage medium for execution by a general purpose computer or a processor. Examples of computer-readable storage mediums include a read only memory (ROM), a random access memory (RAM), a register, cache memory, semiconductor memory devices, magnetic media such as internal hard disks and removable disks, magneto-optical media, and optical media such as CD-ROM disks, and digital versatile disks (DVDs).

Suitable processors include, by way of example, a general purpose processor, a special purpose processor, a conventional processor, a digital signal processor (DSP), a plurality of microprocessors, one or more microprocessors in association with a DSP core, a controller, a microcontroller, Application Specific Integrated Circuits (ASICs), Field Programmable Gate Arrays (FPGAs) circuits, any other type of integrated circuit (IC), and/or a state machine.

A processor in association with software may be used to implement a radio frequency transceiver for use in a wireless transmit receive unit (WTRU), user equipment (UE), terminal, base station, radio network controller (RNC), or any host computer. The WTRU may be used in conjunction with modules, implemented in hardware and/or software, such as a camera, a video camera module, a videophone, a speakerphone, a vibration device, a speaker, a microphone, a television transceiver, a hands free headset, a keyboard, a Bluetooth® module, a frequency modulated (FM) radio unit, a liquid crystal display (LCD) display unit, an organic light-emitting diode (OLED) display unit, a digital music player, a media player, a video game player module, an Internet browser, and/or any wireless local area network (WLAN) or Ultra Wide Band (UWB) module. 

1. In a transceiver, a method for encoding data channels comprising: modulating a non-orthogonal pilot signal using a pilot channel code that is non-orthogonal to codes used to modulate other channels; mixing a data signal with a specific channel code; and combining the modulated non-orthogonal pilot signal and the mixed data signal and generating a composite output signal.
 2. The method of claim 1, further comprising: modulating the combined signal using a PN code.
 3. The method of claim 1, wherein the mixing step comprises modulating the data signal using the specific channel code.
 4. The method of claim 3, wherein the specific channel code is specific to a user.
 5. The method of claim 1, wherein the data signal includes a quadrature and in-phase portion.
 6. The method of claim 6, further comprising: modulating the combined signal using a channel separation signal.
 7. A transceiver for encoding data channels comprising: a pilot spreader modulating a non-orthogonal pilot signal using a pilot channel code that is non-orthogonal to codes used to modulate other channels; a data spreader for modulating a data signal with a specific channel code; wherein the modulated non-orthogonal pilot signal and the modulated data signal are combined to generate a composite output signal.
 8. The transceiver of claim 7, further comprising: modulating the combined signal using a PN code.
 9. The transceiver of claim 8, wherein the specific channel code is specific to a user.
 10. The transceiver of claim 7, wherein the data signal includes a quadrature and in-phase portion.
 11. The transceiver of claim 10, wherein the combined signal is modulated using a channel separation signal. 